High Bandwidth Oscilloscope

ABSTRACT

A method for improving bandwidth of an oscilloscope involves, in preferred embodiments, the use of frequency up-conversion and down-conversion techniques. In an illustrative embodiment the technique involves separating an input signal into a high frequency content and a low frequency content, down-converting the high frequency content in the analog domain so that it may be processed by the oscilloscope&#39;s analog front end, digitizing the low frequency content and the down-converted high frequency content, and forming a digital representation of the received analog signal from the digitized low frequency content and high frequency content.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of i) U.S. Provisional PatentApplication 60/629,050, filed Nov. 18, 2004 by Pupalaikis and entitled“High Bandwidth Oscilloscope,” ii) U.S. Provisional Patent Application60/656,865, filed Feb. 25, 2005 by Pupalaikis et al. and entitled “TheDigital Heterodyning Oscilloscope,” and iii) U.S. Provisional PatentApplication 60/656,616, filed Feb. 25, 2005 by Mueller et al. andentitled “Method and Apparatus for Spurious Tone Reduction in Systems ofMismatched Interleaved Digitizers.” This application is also acontinuation-in-part of U.S. patent application Ser. No. 10/693,188,filed Oct. 24, 2003 by Pupalaikis et al. and entitled “High BandwidthReal Time Oscilloscope,” currently pending, which claims the benefit ofU.S. Provisional Patent Application 60/420,937, filed Oct. 24, 2002 byPupalaikis et al. and entitled “High Bandwidth Real Time Oscilloscope.”

TECHNICAL BACKGROUND

A digital oscilloscope is a tool utilized by engineers to view signalsin electronic circuitry. As circuits and signals get ever faster, it isbeneficial to have digital oscilloscopes capable of digitizing,displaying and analyzing these faster signals. The capability of adigital oscilloscope to digitize fast signals may be measured by itsbandwidth and sample rate. The sample rate is the number of samplespoints taken of a waveform in a given amount of time and is inverselyproportional to the sample period—the time between samples. If asinusoidal frequency sweep is performed from DC up to higherfrequencies, the bandwidth is the frequency at which the signaldisplayed on the digital oscilloscope screen is approximately 30%smaller than the input sine wave.

Since one of the uses of the digital oscilloscope is to design andanalyze new electronic devices, high end digital oscilloscopes generallyoperate at speeds much higher than the present state of the art inelectronics. These speeds may be achieved through the use of ever-fastersampling chips or the use of alternate methodologies to provide thedesired bandwidth.

One such method involves triggering repeatedly on a periodic event. Ifan event is periodically repeating data obtained from multiple triggerevents can be assembled together to provide a good view of the waveform.More particularly, the scope may repeatedly trigger on an event andacquire only a few points of the waveform (sometimes only one point ofthe waveform) on each trigger event. Scopes having this functionalityare sometimes called “sampling scopes.” After repeated triggers, thepoints are reassembled according to the sampling algorithm to create ahigher “effective” sample rate version of the waveform. Furthermore, therepeated trigger events permit averaging, which can be utilized toincrease the signal-to-noise ratio (SNR) and therefore enable furtherbandwidth increases. However, such a sampling scope presupposes arepetitive input signal so that the representation of the waveform canbe generated over many triggers.

This technique may be unsuitable where the signal that is to be analyzedis not repetitive. For instance, the user of the oscilloscope may wantto capture a non-repetitive event such as the cause of some failure inan electronic system. The trigger event may happen repeatedly but thesignal around the trigger event may be different. Therefore, it isdesirable to achieve a high bandwidth and sample rate with only a singletrigger event. Such digital oscilloscopes are sometimes called real-timescopes, and acquisitions taken utilizing only a single trigger event arecalled single-shot acquisitions.

In real-time digital oscilloscope design the required sample rate of thesampling system is a function of the bandwidth of the analog signal tobe acquired. In order to accurately represent the signal the sample rateof the sampling system should be at least twice that of the highestfrequency being digitized. This is often called the “Nyquist rate.”

One method for improving sample rate is time interleaving. This methodutilizes multiple digitizing elements that sample the same waveform atdifferent points in time such that the waveform resulting from combiningthe waveforms acquired on these multiple digitizers forms a high samplerate acquisition. For example, in a system having a twoanalog-to-digital converters, or ADCs, the first ADC samples the signal,then the second ADC samples the signal, then the first and so on. Thedigital output of the ADCs may then be multiplexed or otherwise combinedto yield a composite digital corresponding to the analog input signal.Use of interleaving accordingly eases the speed requirements of each ofthe individual ADCs.

Use of interleaving in digital oscilloscopes may accordingly provide thesignificant advantage of increasing the effective bandwidth of theoscilloscope. With a given set of ADCs, a substantially higher samplerate may be achieved with the use of interleaving. Increasing the samplerate correspondingly increases the maximum frequency that may be sampledby the system, which is commonly called the “bandwidth” of theoscilloscope. The term bandwidth actually refers to a frequency rangerather than an upper limit. The lower end of the range is generallyunderstood to be around 0 Hz for an oscilloscope, so the nominalbandwidth of an oscilloscope generally corresponds to the maximumfrequency that can be sampled by the system. Thus, a two-fold increasein sample rate can provide around a two-fold increase in oscilloscopebandwidth.

Where interleaving is employed the timing relationship, gain, and offsetof each digitizing element is usually matched. When digitizers aremismatched in these characteristics the accuracy of the digitizedwaveform is compromised.

One symptom of mismatched digitizers is error signal generation. Aspecific type of error signal is an artifact signal created by errors inthe interleaving process. One common artifact signal is a spurious tone.When multiple digitizers work in an interleaved configuration todigitize a waveform and a single tone is applied to the system, multipletones result. The frequency location of the spurious tones is determinedby the input frequency and the number of digitizers employed. Themagnitude and phase of the spurious tones is determined by the inputfrequency magnitude and phase, as well as the response characteristicsof the individual digitizers, including the response characteristics ofthe various signal paths leading to each digitizing element. Thesespurious tones serve to degrade the quality of the digitizing system, asmeasured with the aforementioned specifications.

These and other design issues impose practical restrictions on thedegree or order of interleaving used in digital oscilloscopes. Furtherimprovement of bandwidth in digital oscilloscopes has generally beenaccomplished by design and use of faster front-end amplifiers and ADCs.The performance of the amplifiers and samplers, however, is generallylimited by the state of the art in integrated circuit fabrication.

SUMMARY

A method for improving bandwidth of an oscilloscope involves, inpreferred embodiments, use of frequency up-conversion anddown-conversion techniques to achieve a system bandwidth thatsignificantly exceeds the bandwidth attainable through interleavingalone. In an illustrative embodiment the technique involves separatingan input signal into a high frequency content and a low frequencycontent, down-converting the high frequency content in the analog domainso that it may be processed by the oscilloscope's analog front end,digitizing the low frequency content and the down-converted highfrequency content, and forming a digital representation of the receivedanalog signal from the digitized low frequency content and highfrequency content. The digital representation is formed in variousimplementations by up-converting the content to its original frequencyband and then recombining the high frequency content with the lowfrequency content. In preferred implementations this is achieved bypassing the high frequency content through a high pass filter, mixingthat content with a sinusoidal waveform to generate higher and lowerfrequency images, substantially eliminating the higher frequency imagewith a low pass filter, digitizing and upsampling the lower frequencyimage, mixing the digitized and upsampled content with a periodicwaveform having substantially the same frequency as the sinusoidalwaveform to generate higher and lower frequency images, and thencombining the higher frequency image with the low frequency content toform a digital representation of the original input waveform. In thismanner preferred embodiments achieve the significant advantage thatbandwidth may be enhanced beyond the limits associated with interleavingand the state of the art in amplifier and ADC design.

Also disclosed herein is an artifact signal correction system that inpreferred embodiments is used to compensate for error tones generatedduring interleaving. The artifact signal correction system may include amixing component to generate a waveform corresponding to an artifactsuch as an error tone, whereupon that waveform may be combined with theinput waveform to substantially eliminate the artifact. In oneembodiment, an input waveform and a periodic digital waveform are fedinto a mixer to generate a mixed waveform with substantially the samefrequency content as the input waveform except that the frequency isreversed and the phase content is negative. The periodic digitalwaveform may be synchronized to the digitizing elements such that thewaveform has a positive magnitude during portions of the waveformsampled by a first digitizing element and a negative magnitude duringportions of the waveform sampled by a second digitizing element. Themixed waveform may then be input to a digital filter that converts thephase and amplitude of the error tone to substantially the same phaseand amplitude of the corresponding tones in input waveform. Theconverted and mixed waveform may then be synchronized with the inputwaveform by applying a delay to the input waveform that accommodates theaforementioned mixing and converting operations. An inverted version ofthe mixed waveform may then be added to the input waveform so as tosubstantially reduce or eliminate the error tones.

Phase differences between multiple frequency bands may be accommodatedin preferred embodiments by providing a signal processing system thatcompensates for the relative phase difference so that the combination ofthe bands is constructive throughout a substantial portion of a bandoverlap or crossover region. In one embodiment, a signal combiningsystem may include a comparator for determining a relative phasedifference between the two signals within a predefined crossover region,a phase adjusting element for adjusting a phase of one of the twosignals, and a combiner for combining the phase-adjusted signal with theother of the two signals. In another aspect, a method for adjusting aphase relationship between signals from multiple frequency bands thatare being summed may include filtering a first of the signals byapplying an integer samples delay, a fractional sample delay filter, andan allpass filter bank; and summing the filtered first signal with asecond signal.

The details of various additional features are set forth in theaccompanying drawings and the description below. Other aspects andadvantages will become apparent from the description, the drawings, andthe claims.

BRIEF DESCRIPTION OF DRAWINGS

For a more complete understanding of the preferred embodiments,reference is made to the following description and accompanyingdrawings, in which:

FIG. 1 is a schematic representation of an oscilloscope;

FIG. 2 is a block diagram representation of a two channel oscilloscopefront end;

FIG. 3 is a graphical representation of radio frequency (RF) power ateach stage of a high frequency (HF) signal path;

FIG. 4 is a graphical representation of an overall gain at each stage ofthe HF signal path;

FIG. 5 is a graphical representation of a noise power at each stage ofthe HF signal path;

FIG. 6 is a graphical representation of a signal-to-noise ratio (SNR) ateach stage of the HF signal path;

FIG. 7 is a graphical representation of an overall noise metric at eachstage of the HF signal path;

FIG. 8 is a block diagram representation of a signal processingconfiguration;

FIG. 9 is a block diagram representation of a digital signal processing(DSP) system;

FIG. 10 is a representation of a configuration menu;

FIG. 11 is a block diagram representation of a calculation of the phaseof a reference tone;

FIG. 12 is a graphical representation of a low frequency (LF) low pass(LP) filter magnitude response;

FIG. 13 is a graphical representation of a high frequency (HF) low imagefilter magnitude response;

FIG. 14 is a graphical representation of a HF notch filter magnituderesponse;

FIG. 15 is a graphical representation of the combination of the HF lowimage and the notch filter response;

FIG. 16 is a graphical representation of the combination of the HF lowimage and notch filter response showing rejection at the reference tonefrequency;

FIG. 17 is a representation of a digital local oscillator (LO) tonegenerator;

FIG. 18 is a graphical representation of the digitally mixed combinationof the HF low image and the notch filter response;

FIG. 19 is a graphical representation of the digitally mixed combinationof the HF low image and the notch filter;

FIG. 20 is a graphical representation of the HF high image filtermagnitude response;

FIG. 21 is a graphical representation of the overall HF digital filterresponse;

FIG. 22 is a graphical representation of a LF and HF path digital filterresponse;

FIG. 23 is a graphical representation of a LF and HF path digital filterresponse;

FIG. 24 is a graphical representation of a LF and HF path digital filterresponse;

FIG. 25 is a digital oscilloscope screen showing a horizontal settingsmenu;

FIG. 26 is a digital oscilloscope screen fragment showing an internalacquisition configuration; and

FIG. 27 is a digital oscilloscope screen fragment showing acquisitionsystem settings.

Like reference numbers and designations in the various drawings indicatelike elements.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The following description details certain operational and structuralaspects of various preferred implementations and discusses relateddesign considerations. These implementation details are intended to beillustrative and not limiting. Unless expressly stated otherwise, itshould be understood that the components and design approaches describedbelow may be modified to suit other particular applications.

FIG. 1 shows a digital oscilloscope [11] oscilloscope having four inputchannels CH1 [1], CH2 [2], CH3 [3], and CH4 [4] connected to the inputsto each of four front-ends [5] [6] [7] [8]. In accordance with theembodiments described herein, the oscilloscope [11] may be configured todigitize waveforms at sample rates of up to 20 GS/s at bandwidths up to6 GHz into memories up to 50 Mpoints long. Channels 1 [1] and 2 [2] aregrouped together into a channel 1/2 grouping [9] and channels 3 [3] and4 [4] are grouped together in a channel 3/4 grouping [10]. In thechannel 1/2 grouping [9], the channel 1 input [1] is connected to an RFrelay [12] and the channel 2 input [2] is connected to a diplexer [14].The low frequency band (DC-6 GHz) diplexer output [16] in the channel1/2 grouping [9] is connected to oscilloscope front-end 2 [6].Similarly, for the channel 3/4 grouping [10] the low frequency diplexeroutput [17] is connected to front-end 3 [7]. In the normal relaysetting, the channel 1/2 grouping [9] RF relay [12] connects the inputchannel 1 [1] to oscilloscope front end 1 [5] and the channel 3/4grouping [10] RF relay [13] connects the input channel 4 [4] to theoscilloscope front-end 4 [8].

In a high bandwidth mode of operation, the oscilloscope channel 2 [2]utilized for the low frequency band continues to deliver the frequencyband of the input signal from DC to 6 GHz from the diplexer output [16]to oscilloscope front-end 2 [6] and oscilloscope channel 3 [3] deliversthe diplexer output [17] to front-end 3 [7]. In this mode, however, thehigh frequency band (from 6 to 11 GHz) output from the diplexer [18]enters the channel [20](depicted simply as a mixer symbol). The channelserves to translate this frequency band down to 500 MHz to 5.5 GHz. Thistranslated output at the channel output [22] is connected to the RFrelay [12] which connects this signal to oscilloscope front-end 1 [5]Similarly, the high frequency band output from the diplexer [15] highfrequency band output [19] enters a channel [21] and its output [23] isconnected to oscilloscope front-end 4 [8] through RF relay [13]. In thismanner, oscilloscope front-ends 1 [5] and 2 [6] are receiving twofrequency bands. The band received by oscilloscope front-end 1 [5] isdesignated as the HF band because it is receiving the high frequencycontent of the signal applied to the channel 2 input [2]. The bandreceived by oscilloscope front-end 2 [6] is designated as the LF bandbecause it is receiving the low frequency content of the signal appliedto the channel 2 input [2]. Similarly, oscilloscope front-ends 3 [7] and4 [8] are receiving the LF and HF bands, respectively, of the signalapplied to the channel 3 input [3].

The oscilloscope acquires the LF and HF frequency bands simultaneouslyand during waveform processing, translates the HF band back to itsproper location at 6 to 11 GHz, after which the LF and HF bands arerecombined forming an 11 GHz bandwidth acquisition. The result of thisprocessing is two waveforms, one produced from the channel 2 input [2]and the other from the channel 3 input [3]. Channel inputs 1 [1] and 4[4] are disabled for this operation.

During this processing, the digitizing elements of both channels areused resulting in a doubling of sample rate and since both channels areutilized, a doubling of the memory length occurs also. The result is anear doubling of the bandwidth.

While FIG. 1 shows a two channel implementation, any number of channelsmay be employed. In this exemplary implementation, channels 1 [1] and 2[2] are grouped together as a channel 1/2 grouping [9] and channels 3[3] and 4 [4] are grouped together in a channel 3/4 grouping [10].

As noted above, when channel 2 is selected for high bandwidth modeoperation channel 1 [1] is disabled and front-ends 1 [5] and 2 [6] areutilized together to perform high bandwidth acquisitions of the inputapplied to channel 2 [2]. So, when high bandwidth mode is selected forchannel 2, this configuration is referred to as high bandwidth modechannel 2.

FIG. 25 illustrates this grouping and selection from a user interface(UI) perspective. FIG. 25 shows an oscilloscope screen with thehorizontal configuration menu [137] shown. In this menu, oneconfiguration block is the high bandwidth configuration [138]. The highbandwidth configuration [138] shows the two channel groupings: channels1/2 [139] and channels 3/4 [140]. Each grouping has two buttons thattoggle between normal operation of the channels 1/2 [141] and channels3/4 [142] and high bandwidth mode channel 2 [143] and high bandwidthmode channel 3 [144]. FIG. 25 happens to show the oscilloscope operatingin high bandwidth mode on high bandwidth channels 2 and 3.

Considering high bandwidth channel 2 operating in high bandwidth mode,the DC to 6 GHz band is provided to oscilloscope front-end 2. This bandis designated as the low-frequency band or LF band and the path from thechannel 2 input through the system is referred to as the LF path. Theband from 6 to 11 GHz output from the diplexer passes through the highbandwidth circuitry and is eventually connected to oscilloscopefront-end 1 through the RF relay. This band is referred to as the HFband and its path through the system is called the HF path.

While this embodiment has two frequency bands per high bandwidthchannel, optionally three, four or more frequency bands may be used perhigh bandwidth channel. It is advantageous to configure the bands suchthat they are adjacent, as will be appreciated from the descriptionbelow, although nonadjacent bands may be utilized in certainapplications.

FIG. 2 shows a block diagram depicting the high bandwidth hardware. Theblock diagram shows two high bandwidth channels with high bandwidthchannel 2 [27] shown at the top and high bandwidth channel 3 [28] shownat the bottom, with shared circuitry in between. In FIG. 2, the channel2 input [27] enters a diplexer [29] which splits the signal into twofrequency bands. The diplexer preferably provides good voltage standingwave ratio (VSWR) performance. Said differently, the diplexer preferablyreflects as little of the input signal as possible. The DC to 6 GHz band[31] exits the diplexer [29] and passes through a notch filter [30]designed to specially notch out any 10 GHz in the LF path that canemanate from the oscilloscope front end 2 [32] due to a 10 GHz sampleclock used internal to the oscilloscope. The LF signal then continuesdirectly to oscilloscope front end 2 [32]. The remainder of the LFsignal path includes the oscilloscope front-end and 20 GS/s digitizerassociated with oscilloscope front-end 2 [32].

The HF signal path will now be described in greater detail. The HF bandfrom 6 to 11 GHz [33], upon exiting the diplexer [29], enters anattenuator [34]. This attenuator [34] is designed to apply 9 dB ofattenuation at lower frequencies and an extra 6 dB of attenuation (for atotal of 15 dB attenuation) at higher frequencies. The breakpoint forthis frequency dependent attenuator is around 6 GHz. The reason for thisfrequency dependent attenuation is due to the fact that the diplexer[29] pass band edge frequencies have been designed to provide the fullDC to 6 GHz on the LF path, and therefore there is some attenuation ofthe 6 to 11 GHz band at lower frequencies. This frequency dependentattenuator puts back some of the signal content at low frequenciesrelative to high frequencies, thus serving to flatten the HF band. TheHF band signal then enters a variable attenuator [35], designed toprovide from 1 to 41 dB of attenuation. The HF band signal continuesinto a variable gain amplifier (VGA) [36] designed to provide from 13 to24 dB of gain. Both the variable attenuator [35] and VGA [36] aredesigned to provide an overall amount of gain/attenuation to fix the VGAmaximum output power at a constant level corresponding to a full-scalesignal on the screen of the digital oscilloscope at the Volts/div (vdiv)setting specified by the oscilloscope user. The VGA maximum output levelutilized is designed to be −10.48 dBm. The implication of this levelwill be described later.

The output of the VGA [36] passes to another band pass filter [37],through a 3 dB pad [38] and into the radio frequency (RF) input [39] ofa mixer [40] at a maximum level of −14.48 dBm. The implication of thislevel will be described later. Since high frequency mixers tend not tofully isolate the RF input [39] and the intermediate frequency (IF)output [42], the band pass filter [37] is provided to sharply limit thefrequency content to between 6 and 11 GHz. The RF-IF isolation of themixer is typically −20 dB. This means that any signal entering the mixerRF input [39] can appear at the IF output [42] at the same frequency,but attenuated by only 20 dB. Furthermore, because one of the primaryspecifications of the diplexer [29] is input return loss (or good VSWR,as described earlier), some of the frequency rejection of the diplexer[29] has been traded off for better return loss and this band passfilter [37] is designed to supply the extra rejection.

The mixer local oscillator (LO) input [41] originates from anYttrium-Iron-Garnet (YIG) tuned, fixed frequency, internallyphase-locked oscillator (PLO) [43]. The PLO [43] multiplies an internal100 MHz reference by 115 to provide an 11.5 GHz output [44] at an outputpower of 21 dBm. The 100 MHz reference is provided at an output [45] ofthe PLO and is connected to the oscilloscope 100 MHz reference input(not shown). This is to provide a fixed frequency and phase relationshipbetween the LO of the high bandwidth unit and the sample clock of theoscilloscope. The 11.5 GHz output [44] passes through a 20 dBdirectional coupler [46] that picks off the LO for input to a frequencydivider [47] whose purpose will be explained later. Then, the signal issplit through a Wilkinson divider [48] whose purpose is to deliver the11.5 GHz LO to both high bandwidth channels through an 11.5 GHz bandpass filter [49]. The band pass filter [49] is utilized because themixers may not have adequate RF-IF isolation, RF-LO isolation and/orIF-LO isolation. The band pass filter [49] limits the crosstalk betweenthe two high bandwidth channels that share a common LO.

The LO enters the mixer LO input [41] at around +15 dBm and mixes withthe 6 to 11 GHz band present at the mixer RF input [39]. The result atthe mixer IF output [42] is two images of the 6 to 11 GHz band atdifferent frequencies. The low frequency image appears in a band from500 MHz to 5.5 GHz and is reversed in frequency. In other words, 6 GHzat the mixer RF input appears at 5.5 GHz at the IF output and 11 GHz atthe mixer RF input appears at 500 MHz at the IF output. The highfrequency image appears in a band from 17.5 GHz and 22.5 GHz. Inaddition to these two bands, because of imperfect LO-IF isolation, thereis also an amount of 11.5 GHz leakage from the LO input [41] present atthe mixer IF output [42]. The LO leakage power is about 0 dBm. Thedesired signal power in the 500 MHz to 5.5 GHz band is at about −23 dBm.

The mixer IF output [42] passes through a 3 dB pad [50] and enters animage reject low pass filter [51] designed together with a notch filterthat rejects 5.75 GHz. The purpose of the notch filter will be describedbelow. The main purpose of the filter [51] is to substantially rejectthe high frequency image along with the 11.5 GHz LO leakage.

The output of the low pass filter [51] passes to a fixed gain amplifier(FGA) [52] that amplifies the signal to approximately 2 dB maximumoutput power. The implication of this power level will be describedbelow.

The FGA [52] output enters a DC reject filter [53] and enters a 6 dBpower splitter/combiner [54] whose purpose is to combine the HF bandpresent at one input [55] with an LO reference pilot tone present at theother input [56]. In an alternate embodiment a summing amplifier may beused in lieu of splitter/combiner [54].

The generation of the LO reference pilot tone will now be described. Asmentioned previously, the 11.5 GHz LO [44] is picked off through a 20 dBdirectional coupler [46] and exits the pickoff at around 1 dBm. This 1dBm LO signal enters a frequency divider [47] whose purpose is to dividethe 11.5 GHz signal frequency in half thereby producing a 5.75 GHzoutput. The frequency divider has differential outputs. Therefore, itcan be viewed as producing two single ended outputs [57] and [58] thatare out of phase with each other by 180 degrees.

This 5.75 GHz signal, called the LO reference pilot tone, or simply theLO reference passes through DC blocking capacitors [59] and through aband pass filter [60] and through an attenuator [64] such that it isthen attenuated to approximately 50 mV pk-pk, or about −22 dBm prior toentering the splitter/combiner [54]. The purpose of the band pass filter[60] is to pass the LO reference while limiting the crosstalk betweeneach high bandwidth channel due to the mutual use of the singlefrequency divider.

The splitter/combiner output [61] consists of the sum of a frequencyband in the 500 MHz to 5.5 GHz range (representing the heterodynedversion of the band from 6 to 11 GHz, reversed in frequency) and the5.75 GHz LO reference. The reference tone is at −28 dBm and the maximumHF signal power is at −4 dBm. Said differently, the HF band representsthe signal content in the 6 to 11 GHz band, downconverted and frequencyflipped to a band from 500 MHz to 5.5 GHz, with a small 5.75 GHz toneriding on the signal.

As mentioned previously, the RF relay [62] serves to switch the input tofront-end 1 [63] between the oscilloscope input channel 1 [1] and thehigh bandwidth circuitry output of the splitter/combiner [54], so inhigh bandwidth mode, the relay is connecting this HF band tooscilloscope front-end 1 [63]. The remainder of the HF signal pathincludes the oscilloscope front-end and 20 GS/s digitizer associatedwith oscilloscope front-end 1 [63].

Before discussing how the waveforms are acquired and processed fromfront-ends 1 [63] and 2 [32] to form the high bandwidth channel 2waveform, it is useful to discuss some considerations in the design ofthe high bandwidth circuitry. One consideration is dynamic range. Theoscilloscope [11] allows seven different fixed Volts/division (vdiv)settings. These are 10, 20, 50, 100, 200, 500 mV/div and 1 V/div. Theoscilloscope screen has 8 vertical divisions, and as seen in Table 2,this amounts to full-scale signals ranging from −18 to +22 dBm, or about40 dB of dynamic range. As mentioned previously, the high bandwidthcircuitry consists of a variable attenuator [35] and VGA [36], whosecombination of attenuation and gain serves to provide a constant maximumpower at the VGA [36] output of −11.48 dBm. This is accomplished throughthe use of attenuator and gain settings as specified for each vdiv inTable 2. The attenuations are set by digital control of the variableattenuator [35] and the variable gain is set by digital control of adigital-to-analog converter (DAC). The DAC provides an analog levelcorresponding to an input code and the analog level supplied to the VGA[36] controls its gain from between 13 and 24 dB. Table 1 shows adescription and stage number assigned to each stage of the HF signalpath, along with the gain/attenuation of each stage along with the noisefigure.

TABLE 1 Stage # Stage Description Gain/Attenuation Noise Figure 0 Input0 0 1 Diplexer −1 1 2 Fixed Attenuator −8 8 3 Frequency Dependent −6 6Attenuator 4 Variable Attenuator −10 · atten_(vd) − 1 10 · atten_(vd) +1 5 Variable Gain Amplifier 13 + gain_(vd) 1.9 (VGA) 6 BandPass Filter−1 1 7 Pad −3 3 8 Mixer −8.5 9 9 Pad −3 3 10 LowPass Filter −0 0 11Fixed Gain Amplifier 28 3 (FGA) 12 Splitter/Combiner −6 6

It should be noted that the gains for stages 4 and 5 are variable andare determined by Table 2. Note that only a subset of the stages areshown.

TABLE 2 Variable Variable FullScale Input Power Attenuator Gain SettingVolts/Division (mV) (dBm) Setting (dB) 10 −17.96 0 10.48 20 −11.94 04.46 50 −3.98 1 6.5 100 2.04 1 0.48 200 8.06 2 4.46 500 16.02 3 6.5 100022.04 3 0.48

The result of the settings in Table 2 sets the maximum power at allstages at and downstream from the VGA [36] output (stage 5). This can beseen in FIG. 3, where the maximum signal power is seen at each stage.The input power in FIG. 3 varies between −17 and +22 dBm depending onthe oscilloscope vdiv setting, but the stage power is constant from theoutput of stage 5 onward. Thus it is seen that the system design plusthe gain/attenuation settings provide for the desired input range.

Another consideration is noise. The oscilloscope referred to in theembodiment has a signal-to-noise ratio (SNR) of approximately 36 dB.This may be calculated by considering the full-scale range of theoscilloscope at a given vdiv setting relative to the amount of noisepresent. If the noise is assumed to be “white noise”, or in other words,is assumed that the noise power is equally spread over the 6 GHzbandwidth of the instrument, it can be assumed that a doubling ofbandwidth would produce a 3 dB decrease in SNR to 33 dB best case,considering the fact that even if the high bandwidth circuitry isnoise-free, the oscilloscope channel serves to limit the maximumachievable SNR. Since the high bandwidth circuitry is not typicallynoise-free, the design goal is to minimize the degradation of SNR belowthe 33 dB theoretical best case. It has been calculated that a 1 dBdegradation of SNR can be tolerated and that a 32 dB SNR can be achievedif the HF path can be kept above approximately 39 dB.

To calculate this, first, the input noise power may be calculated usingthe theoretical noise power of a 50 Ohm resistor at room temperature as−173.91 dBm/Hz or −76.13 dBm in a 6 GHz bandwidth. This is the inputnoise power as shown in FIG. 5. Then, the cascaded noise figure may becalculated using accepted techniques for each of the vdiv settingsspecified, along with the gain/attenuation settings for these vdivs andnoise figures of each stage as shown in Table 1. Suitable cascaded noisefigure calculations are described in Adam, Stephen F., Microwave Theoryand Applications, Prentice Hall, 1969, pp 490-500. The calculatedoverall noise figure at each stage in the HF signal path is shown inFIG. 7 where only the final stage noise figure is particularlysignificant. This noise figure may then utilized to calculate the noisepower at each stage in FIG. 5. The stage power shown in FIG. 3 alongwith the noise power shown in FIG. 5 may be used to calculate the SNR ateach stage, as shown in FIG. 6. FIG. 6 shows that the final SNR of theHF signal path is kept above 39 dB for all vdiv cases. A similar seriesof calculations can be made for any desired SNR specification.

Another consideration is that of distortion. Distortion can produceunwanted tones, or spurs, in the frequency spectrum and serves todecrease the effective number of bits (ENOB) of the system. This isbecause the ENOB calculation includes noise and distortion componentsand not just the noise. Also, the size of the largest spur determinesthe spur-free dynamic range (SFDR), another relevant specification. Itshould be noted that spurs may be created inside the oscilloscopefront-end and digitizer due to the waveform digitizing process. It maybe deemed acceptable if no spurs significantly degrade the originaloscilloscope performance. As an example, one design goal may be to keepall spurs created by the high bandwidth hardware below −40 dBc.

Distortion may be created by the two active components, the FGA [52] andVGA [36], along with the mixer [40]. Distortion components are kept to aminimum generally by good amplifier and mixer design, and by keepingpower levels low in these devices.

In the case of the mixer [40], one factor affecting distortion is themixer topology. In the preferred implementation, a triple-balanced,“medium level” mixer is utilized. The term “medium level” refers to thepower of the LO delivered to the mixer LO input [41]. A second factor isthe method of downconversion. The frequency downconversion process canbe implemented with the LO frequency placed at a frequency below thefrequency band of interest (low side downconversion) or at a frequencyabove the band of interest (high side downconversion). Low sidedownconversion preserves the frequency order of the band (i.e., it doesnot flip the frequency) but tends to produce a significant number ofrelatively large spurs. High side downconversion, while creating thecomplication of flipping the frequencies in the band, produces less andsmaller spurs. The preferred implementation utilizes high bandwidthdesign utilizes high side downconversion although either method issuitable.

Table 3 shows certain mixer specifications as they pertain to distortioncomponents. The mixer utilizes a 15 dBm LO power. Typically, the inputthird order intercept point (IIP3) is 5 dB above the LO power and theinput second order intercept point (IIP2) is 10 dB above that. Aspreviously mentioned, the mixer RF input power is in this implementationkept at or below −14.48 dBm. Table 3 also shows the calculation of thefour largest distortion products: the second and third orderintermodulation products due to two tones (IM2 and IM3, respectively)and the second and third harmonics (H2 and H3). These have beencalculated using commonly used rules-of-thumb supplied in Genesys 2004RF Microwave Design Software—Simulation, Eagleware Corporation, 2004, pp108-110. These tend to be somewhat conservative calculations and theactual distortion components may be substantially less. Table 3 showsthat the largest calculated distortion component is IM2 at −44.48 dBc.

TABLE 3 Power Specifications Levels Definition Specification GovernanceLO Power 15 dBm P_(LO) By Selection of LO Generator plus System DesignInput 3^(rd) Order 20 dBm Typically IIP3 = P_(LO) + 5 By Mixer DesignIntercept Point (IIP3) Input 2^(nd) Order 30 dBm Typically IIP2 = IIP3 +10 By Mixer Design Intercept Point (IIP2) RF input power −14.48 dBm   P_(RF) By Gain Stackup in System Design Distortion Components LevelsPower Level Calculation Frequency Locations Second order −44.48 dBc IM2= −(IIP2 − P_(RF)) F1 + F2, F1 − F2 intermods (IM2) Third orderintermods −68.96 dBc IM32 = −2 · (IIP3 − P_(RF)) 2 · F1 − F2, 2 · F1 +F2, due to two tones (IM32) 2 · F2 − F1, 2 · F2 − F1 2^(nd) Harmonic(H2) −50.48 dBc H2 = IM2 − 6 2 · F 3^(rd) Harmonic (H3)  −78.5 dBc H3 =IM32 − 9.542 3 · F

In the case of the VGA [36], one distortion-related consideration is theoutput 1 dB compression point (P1dB). This is the point where the outputis compressed to the point that an output signal appears 1 dB lower.This point typically occurs near the saturation level of the amplifier.To avoid distortion, it may be desirable to keep the output power wellbelow this point. Table 4 shows the VGA specifications and distortioncalculations. These calculations are similar to that of the mixercalculations. Table 4 shows that with the aforementioned limit on VGAoutput power at −10.48 dBm, the largest distortion components are IM2 at−40.48 dBm and H2 at −46.48 dBm. Again, these tend to be somewhatconservative calculations and the actual distortion components may besubstantially less.

TABLE 4 Power Specifications Levels Definition Specification Governance1 dB compression 10 dBm P1dB By Amplifier Design point (P1dB) Output3^(rd) Order 20 dBm Typically By Amplifier Design Intercept Point (OIP3)OIP3 = P1dB + 10 Output 2^(nd) Order 30 dBm Typically By AmplifierDesign Intercept Point (OIP2) OIP2 = OIP3 + 10 Output power −10.48dBm    P_(VGA) By Gain Stackup in System Design Distortion ComponentsLevels Power Level Calculation Frequency Locations Second order −40.48dBc IM2 = −(IIP2 − P_(VGA)) F1 + F2, F1 − F2 intermods (IM2) Third orderintermods −60.96 dBc IM32 = −2 · (IIP3 − P_(VGA)) 2 · F1 − F2, 2 · F1 +F2, due to two tones (IM32) 2 · F2 − F1, 2 · F2 − F1 2^(nd) Harmonic(H2) −46.48 dBc H2 = IM2 − 6 2 · F 3^(rd) Harmonic (H3)  −70.5 dBc H3 =IM32 − 9.542 3 · F

The FGA [52] calculation is similar to the VGA calculation. Because theFGA output is set at +2 dBm, a higher power amplifier is utilized with aP1 dB of +20 dBm, as shown in Table 5. Table 5 shows that the largestdistortion components are IM2 at −38 dBm and H2 at −44 dBm. Again, thecalculations tend to be conservative and the actual distortioncomponents may be smaller.

TABLE 5 Power Specifications Levels Definition Specification Governance1 dB compression 20 dBm P1dB By Amplifier Design point (P1dB) Output3^(rd) Order 30 dBm Typically By Amplifier Design Intercept Point (OIP3)OIP3 = P1dB + 10 Output 2^(nd) Order 40 dBm Typically By AmplifierDesign Intercept Point (OIP2) OIP2 = OIP3 + 10 Output power 2.021 dBm  P_(FGA) By Gain Stackup in System Design Distortion Components LevelsPower Level Calculation Frequency Locations Second order −37.98 dBc IM2= −(IIP2 − P_(FGA)) F1 + F2, F1 − F2 intermods (IM2) Tnird orderintermods −55.96 dBc IM32 = −2 · (IIP3 − P_(FGA)) 2 · F1 − F2, 2 · F1 +F2, due to two tones 2 · F2 − F1, 2 · F2 − F1 (IM32) 2^(nd) Harmonic(H2) −43.98 dBc H2 = IM2 − 6 2 · F 3^(rd) Harmonic (H3)  −65.5 dBc H3 =IM32 − 9.542 3 · F

Accordingly, the above described techniques may optionally be used toaddresses performance and signal fidelity aspects such as dynamic range,noise, distortion, crosstalk and input return loss (or VSWR) from thehardware design perspective.

Once high bandwidth mode has been selected (as shown in FIG. 25), the RFrelay [62] may be configured, the variable gain and attenuation may beselected (based on the channel vdiv setting) and applied to the variableattenuator [35] and VGA [36] and input signals may be continuouslyapplied to the oscilloscope front-ends. In the case of the highbandwidth channel 2 input [27], the LF band may be applied tooscilloscope front-end 2 [32] and the HF band passing through theaforementioned high bandwidth hardware is applied to oscilloscopefront-end 1 [63]. The LF band may optionally have no additional gain orattenuation applied by the high bandwidth hardware and is acquired atthe vdiv specified for the high bandwidth channel. As can be seen inFIG. 3, the final maximum output power of the HF band in this embodimentis −3.976 dBm, which, as can be seen in Table 2 is the full-scale powercorresponding to the 50 mV/div oscilloscope setting. Therefore, the HFband in this embodiment is acquired by the oscilloscope at 50 mV/div.

FIG. 27 shows an internal calibration menu that shows the variable gain(shown as Gem VG) [154] and variable attenuation settings (shown as GemAtt) [155] are set. These settings are arrived at during factorycalibration of the unit.

FIG. 26 shows some additional information that may be used to controlhow waveform acquisitions will proceed. FIG. 26 shows acquisitionminimum duration [148], start time [149], and stop time [150]. These aresettings used to control the minimum acquisition time duration, theextra time needed to the left of the waveform and the extra time neededto the right of the waveform, respectively, as it will be shown on thescreen. This extra time may provide for LO reference recovery, whichwill be further described below, and for waveform digital processing,that consumes points for filter startup to the left of the waveform orpossibly consumes points to the right of the waveform to account forfilter delay.

Once the acquisition has been configured, the oscilloscope may arm theacquisition and acquire LF and HF portions of the input signal. Theremainder of this section describes the digital processing of the LF andHF waveforms and the recombination of the waveforms into a single highbandwidth waveform acquisition in an illustrative embodiment.

FIG. 9 is a block diagram of the digital system utilized to process asingle high bandwidth channel. FIG. 8 shows the channel 2 high bandwidthprocessor [66] and the channel 3 high bandwidth processor [67] in theprocessing system (also called the processing web) of the oscilloscope,and FIG. 9 serves as a block diagram showing the operation of one suchprocessor.

The signals applied at the oscilloscope front-ends [32 and 63] areacquired. In the preferred implementation, the acquisition systemincludes interleaved ADCs, buffers, and high speed memory such as SRAMor modified DRAM [not shown] to receive the digitized data stream.

After the waveforms are acquired by the oscilloscope, they may beapplied appropriately to the HF input [70] and LF input [69] in FIG. 9.Each waveform acquired by the oscilloscope may contain not only thewaveform data consisting of an array of voltage levels, but also extrainformation that helps in the interpretation of the data points,including horizontal offset, horizontal interval, number of points, ADCsampling phase, vertical offset, and vertical step. In thisimplementation, horizontal offset is the time (relative to theoscilloscope trigger point) associated with the first point of thewaveform. Also in this implementation, horizontal interval is the timebetween each sample point (the reciprocal being the sample rate) and thenumber of points is the number of points in the waveform. The ADCsampling phase describes which of the two interleaved 10 GS/s digitizerssampled the first waveform point (with the understanding that everyother point is taken from every other digitizer). The vertical offset inthis implementation is the voltage associated with code 0, the lowestADC output. Vertical step is the voltage between each code or step inthe ADC output.

FIG. 27 shows, in addition to the variable gain [154] and attenuation[155] settings determined for each vdiv, a calibration called HF delay[156]. In this implementation HF delay is the measured path delay of theHF path relative to the LF path. Described differently, the LF and HFportions of the signal travel through different paths with the HFportion, in this embodiment, traveling through a comparatively longarray of analog processing elements. This path may delay the HF waveformrelative to the LF waveform. The HF delay value [156] is used to correctfor the calculated difference in path propagation times and depends onthe vdiv setting. A negative HF delay means that the HF path should beadvanced to arrive at the proper time. The high bandwidth system mayaccount for the path propagation time differences in hardware or, as inpreferred embodiments, the propagation amount may be measured andaccounted for in the digital system by adding the HF delay to thehorizontal offset of the HF waveform acquired prior to processing.

In the preferred implementation, prior to processing the waveforms thedigital elements shown in FIG. 9 in the HF and LF path are assembledexcept for the elements designated as adaptors ([72] and [79]) andupsampler and fractional delay filters ([73] and [80]). The filters maybe built according to specifications shown in the dialog in FIG. 10.Once these elements are assembled, the system built with these elementsmay be analyzed to account for three possible effects of each filter:the upsample factor, the startup samples, and the delay. The upsamplefactor is in this embodiment the factor by which the waveform samplerate is increased as it passes through a filter element and is generally1 for all of the filters, except the upsampler and fractional delayfilter, where the upsample factor is generally 2. The startup samples inthis embodiment correspond to the time required for the impulse responseto end or die down to an acceptable amount. In the case of the highbandwidth system, the filters are preferably finite impulse response(FIR) filters for simplicity of design and for simplicity in calculatingdelay and startup and are symmetric (to avoid group delay variations).Asymmetric filters and infinite impulse response filters or otherdiscrete-time or continuous-time filters may be better suited to certainapplications. In the case of the symmetric FIR filter, the startup timeis the filter length and the delay (in samples) may be half the filterlength. The analysis of the system paths with filter upsample factors,startup samples, and delay accounted for produces overall equivalentfilters from the standpoint of these three factors for the digitalsignal paths leading from the waveform inputs to the mixing node and thesumming node. Calculation of these equivalent filters leads to adetermination of integer and fractional delay of each path relative tothe other. The integer delay portion is accounted for in the design ofthe adaptors ([72] and [79]), whose primary purpose is to delay thewaveform an appropriate number of integer samples. The fractional delayportion is in this embodiment accounted for in the design of theupsampler ([73] and [80]).

Each upsampler ([73] and [80]) is designed utilizing a polyphase filterarrangement where each filter phase is calculated by sampling a Sin c(Sin x/x) pulse. Simply shifting the Sin c prior to samplingaccomplishes the fractional delay. Suitable fractional delay filters andupsampling filters (sometimes referred to as interpolating filters)designs may be found in Smith, Julius O., MUS420/EE367A Lecture 4A,Interpolated Delay Lines, Ideal Bandlimited Interpolation, andFractional Delay Filter Design, Stanford University.

In the preferred embodiment all of the digital processing elements arebuilt once at inception, except for the adaptors ([72] and [79]) andupsampler and fractional delay filters ([73] and [80]). These arepreferably built on each waveform acquisition to account for variationsin the horizontal waveform information. These processing elements arealso preferably built so that the processed waveforms arrive at thesumming node at the appropriate time.

Consider the LF input [69] in FIG. 9. The path begins with the LF signalentering the LF Interleave correction filter [71]. A description of asuitable filter [71] is described in the U.S. patent application filedNov. 16, 2005 by Mueller et al. and entitled “Method And Apparatus ForArtifact Signal Reduction In Systems Of Mismatched InterleavedDigitizers,” which is incorporated herein by reference. As explained inthe co-pending patent application, this filter may be designed toimprove the digitizer matching of the two interleaved 10 GS/s digitizersthat produce a 20 GS/s sample rate. As such, it serves to reduce thesize of distortion components resulting from inadequate digitizerfrequency response matching.

The LF waveform then enters the LF adaptor [72], which serves to delaythe waveform by an integer number of samples. The waveform then entersthe upsampler and fractional delay filter [73]. This filter [73], asmentioned previously, serves to increase the sample rate from 20 GS/s togenerally 40 GS/s and to provide fractional sample delay of thewaveform. This upsampling is acceptable as the frequency content of theLF input signal has been band limited to 6 GHz by the diplexer at thehigh bandwidth channel input and by limitations of the oscilloscopefront-end in this embodiment. The upsampler [73] may be configured basedon upsampler settings [93] in the dialog shown in FIG. 10. This dialogspecifies the upsample factor [94], the sample distance [95] and anoptimization [96]. The upsample factor [94] is generally set to 2, buthigher upsample factors can be utilized. The sample distance [95] refersto the distance in samples from the input waveform to apply the sin(x)/xinterpolation. Said differently, it is one half the length of eachfilter phase where the number of phases is determined by the upsamplefactor [94].

Referring once again to FIG. 9, the upsampled LF waveform then enters alow pass filter [74]. The response of the exemplary low pass filter isshown in FIG. 12. The filter [74] is preferably built according to thelow pass filter specifications [97] shown in FIG. 10 and using atechnique called frequency sampling, a suitable description of which isprovided in Jong, Methods of Discrete Signal and Systems Analysis,McGraw Hill, 1982, pg. 369. The low pass filter specifications [97]dictate 400 filter coefficients [98], a low cutoff at 0 [99], a highcutoff at 6 GHz [100] and a transition band of 800 MHz [101]. The mainpurpose of this filter [74] in this embodiment is to reject noise andspurs in the LF path beyond 6 GHz.

The LF waveform then enters a crossover phase correction element [75].As will be shown, in this embodiment there is an approximately 200-300MHz wide region where the LF and HF bands interfere. This region isdesignated as the crossover region. It is preferable that thisinterference be constructive in nature. One way to accomplish this is toprovide that the phase of the LF path relative to the HF path isessentially zero while the bands are transitioning. One relevant designcriterion is that sharp filters tend to have extreme phase changes nearthe band edges. The crossover phase correction element [75] maycompensate for this by making the relative phase approximately zerothroughout the crossover region. A description of a suitable crossoverphase correction element [75] is found in the US patent applicationfiled Nov. 16, 2005 by Pupalaikis, et al., entitled “Method of CrossoverRegion Phase Correction When Summing Multiple Frequency Bands”, thecontents thereof being incorporated herein by reference.

The phase corrected and low pass filtered LF band then enters a scalingelement [76] and then the summing node [77]. This operation will bedescribed following the description of the HF path processing.

The HF waveform [70] enters an interleave correction filter [78], anadaptor [79], and an upsampler and fractional delay filter [80] that inthe preferred embodiment works in the same manner as previouslydescribed for the LF path, but with different internal designspecifications depending on the ADC matching of the HF signal path, andthe delay of the HF path.

Still referring to FIG. 9, the HF waveform then enters the HF low imagefilter [81]. This filter may be designed according to the HF low imagefilter specifications [102] shown in FIG. 10 to provide a response asshown in FIG. 13. The filter may be a symmetric FIR filter builtutilizing frequency sampling methods as described above. Exemplaryspecifications provide 400 filter coefficients [103], a low cutoff at300 MHz [104], a high cutoff at 5.5 GHz [105] and a transition band of800 MHz [106]. This filter [81] may be designed to reject all frequencycontent outside of the band of interest from 500 MHz to 5.5 GHz. It mayalso be designed to reject DC, as any DC offset of the channel might beinterpreted as 11.5 GHz frequency content, which would in turn degradethe signal. Also, the filter [81] may be especially designed to rejectnoise and spurs outside of the band of interest.

The HF waveform then enters a 5.75 GHz notch filter [82]. This filtermay be designed to remove the 5.75 GHz LO reference tone riding on thesignal. The purpose of this tone will be described shortly, but in theHF path to the summing node [77], it is preferably rejected. The notchfilter [82] may be designed according to the notch filter specifications[107] shown in FIG. 10 and its response is shown in FIG. 14. The filteris a single biquad filter section designed utilizing a bilineartransformation to convert the following analog prototype filter todigital:

$\begin{matrix}{{H(s)} = \frac{s^{2} + \omega_{0}^{2}}{s^{2} + {\frac{\omega_{0}}{Q} \cdot s} + \omega_{0}^{2}}} & {{Equation}\mspace{14mu} 1}\end{matrix}$

where ω₀=2·π·f₀ and

$Q = \frac{f_{0}}{\Delta \; f}$

and are dictated by the notch filter specifications of f₀ (the notchfilter frequency [108]) and Δf (the notch filter bandwidth [109]). Thebilinear transformation is discussed further in Pupalaikis, BilinearTransformation Made Easy, ICSPAT 2000 proceedings, CMP Publications,2000. Since the filter in this embodiment is infinite impulse response(IIR), there is no filter length per se so filter startup samples [110]are specified to account for filter startup. While processing of the HFlow image filter is described as taking place before the processing ofthe notch filter, their order may be reversed without any adverse effect(a design option that likewise applies to many of the other filtertopologies described herein, as will be understood in the context ofthis detailed description).

The combined response of the HF low image filter [81] and the 5.75 GHznotch filter [82] is shown in FIG. 15 with a zoomed section showing therejection of the 5.75 GHz LO reference shown in FIG. 16. FIG. 16 showsthat the 5.75 GHz LO reference has been attenuated by approximately 50dB and has been essentially removed from the HF waveform in thepreferred embodiment.

The generation of the digital LO [84] begins in this embodiment with thesplit in the HF path [85]. Before the digital LO is generated, the phaseof the LO may first be determined. The LO phase may be determined basedon the LO reference riding on the HF waveform. Referring back to FIG. 2,one can see that the PLO output [44] may be delivered to the mixer LOinput [41] along one signal path, and may be simultaneously picked off,divided down in frequency, and inserted into the HF waveform as the LOreference at the splitter combiner [54]. This LO reference signal has aconstant phase relationship to the LO waveform delivered to the mixer.It is preferable that the LO reference has a constant phase relationshipto the LO. As such, the LO reference tone may carry the phaseinformation required to determine the phase of the LO (with a constantoffset). The constant offset difference between the LO reference and theactual LO may be accounted for through the calibration of HF delay [156]shown in FIG. 27. As mentioned previously, there is a notch filter [51]in the HF hardware path shown in FIG. 2 that rejects any 5.75 GHz comingfrom the input. Therefore, the 5.75 GHz tone is substantially due to the5.75 GHz LO reference inserted at the splitter/combiner [54] in thisembodiment.

One way to generate the digital LO that is phase locked to the LOreference tone is to utilize a digital phase-locked loop (PLL). A lesscomputationally intensive option, used in this embodiment, makes use ofthe fact that the frequency of the LO reference tone relative to theoscilloscope sample clock is relative stable due to the fact that the100 MHz PLO reference output [45] is supplied to the oscilloscope as thereference that generates the oscilloscope's sample clock. Furthermore,because the LO reference in this embodiment is relatively high infrequency (preferably as high as possible for capture by a oscilloscopefront-end with 6 GHz of bandwidth), only a small number of cycles arerequired to accurately determine the phase of the LO reference.

One way to determine the phase of the 5.75 GHz LO reference is to takethe discrete Fourier transform (DFT) of some number of samples of the HFwaveform and pick out the frequency component that occurs at 5.75 GHz.In this embodiment, the phase of this frequency component is the phaseof the LO reference. Since the sample clock generator in theoscilloscope and the LO are generated using the same 100 MHz reference(i.e. the LO and the sample clocked are locked together), there is noambiguity regarding the exact frequency bin in the DFT containing the5.75 GHz component. In other words, even if there were slight errors inthe exact frequency of the LO and therefore the 5.75 GHz LO reference,these slight errors would occur simultaneously in the frequency of thesample clock, and if one assumed that the oscilloscope sample rate wasexactly 20 GS/s, the LO reference would be measured at 5.75 GHz.

Since the DFT and even the fast Fourier transform (FFT) arecomparatively computationally intensive, and because the DFT providesmore information than is actually needed in this embodiment, analternate method for tone detection is preferred. This method is calledthe Goertzel algorithm and is described in Digital Signal ProcessingApplications Using The ADSP-2100 Family, Prentice Hall, 1990, pg. 458.The block diagram of a digital processing element that accomplishes theLO reference phase is shown in FIG. 11. In FIG. 11, the number of pointsutilized (K) and the frequency bin (n) may be determined by the localoscillator and reference specifications [111] shown in FIG. 10. Thespecifications provide in this embodiment that the LO reference is at5.75 GHz [112], to use a maximum of 5000 cycles for LO determination[113], and that the cycles are to be an integer multiple of 23 [114].The cycles multiple makes the number of samples integer and thereforeallows for phase detection without resorting to the technique ofwindowing. The minimum number of LO reference cycles available in agiven waveform is dictated indirectly by the specification of theminimum acquisition duration [148], as specified in FIG. 26.

Once the phase detector [86] has measured the phase of the LO reference(which is preferably performed for every individual waveform acquired),it may be passed to the digital LO generator [87]. A block diagram ofthe LO generator [87] is shown in FIG. 17. FIG. 17 shows that the tonemay be generated utilizing a lookup table [131] utilizing the localoscillator and reference specifications [111] as shown in FIG. 10. Thespecifications provide in this embodiment that the cycles multiple [114]is 23, which means that the sine wave, regardless of phase, will repeatevery 80 samples. Therefore, a table of 80 sine wave values is generatedfor the lookup table [131]. The lookup table may be utilized tocalculate the LO sine wave value for each point k by looking up thevalue at element mod(k,K) to determine the proper LO waveform value atpoint k.

This means that for every waveform point in the HF signal, anaccompanying LO waveform signal value can be determined that is phaselocked relative to the HF signal, and is substantially similar to avalue that would be determined if the analog LO signal applied to themixer LO input [41] shown in FIG. 2 were sampled along with the HFsignal.

Returning to the description of the HF path, and particularly the mixer[83] in FIG. 9, the digitally generated LO may be multiplied with the HFwaveform applied to the mixer. This digital mixing action causes theinput frequency band from 500 MHz to 5.5 GHz to produce two new images,as shown in FIG. 18. The band located from 6 to 11 GHz [135] containsthe desired frequency content provided in the 500 MHz to 5.5 GHz range,but flipped in frequency. The frequency flipping action caused by thehigh bandwidth hardware due to the high-side downconversion has now beenundone and the frequency band has been restored to its correct frequencyband location. Another image is produced from 12 to 17 GHz [136] that isan undesired image in this embodiment. At this point, the rationale forupsampling in this embodiment can be seen—if the HF waveform had notbeen upsampled, the 12 to 17 GHz band [136] would be aliased into a bandfrom 3 to 8 GHz. Upsampling allows this band to have a benign effect inthis embodiment.

The region around 11.5 GHz should be examined to verify that the properrejection of the input DC component has occurred. Referring to FIG. 19,it is seen that the component at 11.5 GHz has been attenuated by atleast 50 dB.

The HF waveform proceeds from the digital mixer [83] to the HF highimage filter [88]. This filter [88] may be built according to the highimage filter specifications [115] shown in FIG. 10. It may be asymmetric FIR built utilizing frequency sampling methods. Thespecifications provide in the preferred embodiment 400 filtercoefficients [116], a low cutoff at 5.8 GHz [117], a high cutoff at 11.5GHz [118] and a transition band of 500 MHz [119]. Its response is shownin FIG. 20. The main purpose of this filter is to reject the imageproduced by the mixing action in the 12 to 17 GHz range [136] shown inFIG. 18.

The combination of all of the filters in the HF path is shown in FIG.21. This represents the response of the digital system of the preferredembodiment to the HF input.

At this point, the processed LF and HF waveforms are almost ready forsumming. The waveforms are preferably scaled before summing. The scalingof the HF waveform depends in this embodiment on the relationship of thehigh bandwidth channel vdiv setting (which is the same as the LFfront-end vdiv setting) and the 50 mV/div range used to acquire the HFwaveform. The HF waveform scaling is calculated:

$\begin{matrix}{{HFGain} = {\frac{LGVdiv}{HFVdiv} \cdot 2}} & {{Equation}\mspace{14mu} 2}\end{matrix}$

HF vdiv is, in this situation, a constant 50 mV/div and the factor of 2accounts for the fact that each frequency band created by the mixingaction is half size. While this factor could have been accounted for bydoubling the size of the digital LO, the foregoing technique ispreferred where processing is performed within the oscilloscopeutilizing integer arithmetic which could cause an overflow.

After scaling the LF waveform using the LF gain element [76] and HFwaveform using the HF gain element [89], the scaled waveforms may becombined by the summer [77] that adds them together. As noted above,phase shifts, amplitude variations, transport delays, phase responses,and other distortion characteristics of the transport paths have beentaken into account in generation of the filter stages. Therefore, thecombination of the signals in the two channels amounts to a summing ofthe compensated waveforms.

Rather than compensate the waveforms at this earlier stage, thecompensation could be performed at the combining stage. Thus, a morecomplicated filter/summing stage could be employed that would performthe same two basic functions: compensating the signals for distortioncharacteristics generated by propagation trough the transmissionchannels, and combining of the signals into a single waveform having theincreased bandwidth.

The overall response of the digital system of the preferred embodimentas a result of this processing is shown in FIG. 22 where the LF and HFpath response is shown along with the combined response. It can be seenthat the digital processing preserves the 11 GHz bandwidth specificationof this embodiment.

FIG. 23 shows the result of the recombination in the crossover region inthis embodiment. It can be seen that middle of the region occurs at6.035 GHz and the response of each filter is approximately the desired−6 dB. The width of the region (defined as the region where the LF andHF band differs in strength from each other by less than 20 dB)encompasses the frequencies between approximately 5.85 GHz and 6.21 GHz,a band of approximately 360 MHz.

FIG. 24 shows the non-flatness caused by the band combination. It isseen that the non-flatness is on the order of +/−0.5 dB. Furtherimprovements may be made to the flatness of the resulting signal basedon minor alterations of the filter specifications.

The result of the processing in FIG. 9 up to this point is to split thesignal into two frequency bands, inject these two bands into twoseparate oscilloscope front-ends, acquire the waveforms, and digitallyprocess the waveform to provide an 11 GHz waveform acquisition. Theanalog processing of these waveforms may lead to magnitude response andgroup delay non-flatness, which causes distortion in the frequencyresponse and time domain response of the system. For this reason,techniques may be utilized to compensate the magnitude response andgroup delay to provide a good overall response. For example a signalprocessing system capable of compensating for a channel responsecharacteristic of an input waveform may comprises input specifications,a filter builder, and a filter. The input specifications may be used tospecify the design of the filter and include channel responsecharacteristics defining the response characteristics of a channel usedto acquire the input waveform, and user specifications for specifying adesired frequency response and a degree of compliance to the desiredfrequency response. The filter builder may generate coefficients for thefilter and outputs final performance specifications. The filter may havea compensation filter generator for generating coefficientscorresponding to a compensation response on the basis of the inverse ofthe channel response characteristics, and a response filter generatorfor generating coefficients corresponding to a combination of an idealresponse and a noise reduction response on the basis of the userspecifications. The filter filters the input waveform and may outputs anoverall response waveform having a desired frequency response. Thefilter may further comprise a filter coefficient cache for storing thecoefficients generated by the filter builder, a compensation filterportion for filtering the input waveform in accordance with thecoefficients stored in the filter coefficient cache corresponding to thecompensation response, and a response filter portion having a responsefilter stage and a noise reduction stage for filtering the compensatedwaveform output from said compensation filter portion that outputs theoverall response waveform. The response filter portion may filter usingthe coefficients stored in the filter coefficient cache corresponding tothe combination of the ideal response and the noise reduction response.

The final system output of the preferred embodiment is shown in FIG. 25,where an applied step with a rise time of 31 ps has been applied. Thehigh bandwidth equipped oscilloscope acquired this waveform with ameasured rise time of 51 ps, for an internal oscilloscope rise time of40.5 ps, a rise time commensurate with an 11 GHz oscilloscope.

Therefore, in illustrative embodiments acquisition of a continuous timerecord wide-bandwidth signal is performed by a system employing multiplenarrower bands, thereby making the best use of a system having physicalcharacteristic limitations, such as bandwidth and sampling ratelimitations. When the portions of the signal acquired by the multiplenarrower bands are digitized, digitally processed, and recombined in amanner to minimize the effects of processing thereon, a digital outputmay be produced that is a substantially similar representation of thewideband input signal over substantially all of the bandwidth of theoriginal signal.

While this detailed description makes reference to an 11 GHzoscilloscope as an example, the techniques described here are equallyapplicable to oscilloscopes or other devices employing signalacquisition, at any bandwidth.

The analog input signal is preferably received from a probe, eitheractive or passive, in communicative contact with a device under test.The probe may optionally be a current probe or a differential probe.

Moreover, as mentioned above any number of component frequency bands maybe derived from the input signal. Three, four, five or more bands may bederived from the input signal. The bands need not be adjacent oroverlapping. In various embodiments, the bands are spaced apart and omita portion of the input signal frequency band.

The down-conversion of the high frequency content need not be to thesame frequency band as any low frequency band and the up-conversion neednot be to the same frequency as any high frequency band. Any desiredamount of frequency translation may be used. For instance, saidtranslation may generate a signal occupying a frequency band overlappingbut not substantially coincident with an adjacent frequency band.

The processing steps performed after acquisition or digitization can beaggregated or separated as desired. In one embodiment, allpost-digitization processing steps are aggregated into a single digitalprocessing element that outputs a digital representation of the analoginput waveform. In other embodiments, subsets of the aforementioneddigital processing steps, such as error tone removal and delay, aregrouped into discrete digital processing elements.

Each of the digital processing elements may be built at inception ordynamically upon each acquisition. Where multiple discrete digitalprocessing elements are used, some may be built at inception and othersmay be re-built each time a signal is acquired, as appropriate.

The down-conversion of the high frequency content can occur in thepreferred embodiments at any point prior to digitization.Pre-digitization filtering and signal processing can be performed on thecomposite analog input signal before the signal is separated or the highfrequency content is otherwise down-converted.

The periodic waveforms used in the frequency conversion techniquesdiscussed herein need not be sinusoidal. Other periodic waveforms suchas impulse trains and square waves may be utilized in certainembodiments.

Dynamic range, noise, distortion, crosstalk and input return loss (orVSWR) compensation are each optional. In various embodiments some or allof these compensations are omitted.

The digital representation of the analog input signal formed byrecombining the two frequency bands may be output to downstream digitalsignal processing hardware to further filter or otherwise process thesignal.

It will be understood by those skilled in the art that a signal ofinterest may not span the entire bandwidth of the analog input signal.In preferred embodiments, the entire input bandwidth is processed anddigitized in the manner described even if a signal of interest (such aclock signal of a system under test) occupies only a small fraction or apoint of the system bandwidth.

Unless expressly stated otherwise the particular circuit topologiesdescribed herein are merely illustrative architectures suitable forparticular preferred embodiments. For instance, the input signal neednot diplexed. Other approaches, such as passing the signal through a 50ohm splitter and then through a bandpass filter may be utilized instead.Similarly, mixer isolation issues can be rejected with an image rejectmixer or triple-balanced mixers. The notch filtering may be omittedwhere a different master clock is utilized. The LF content may beamplified or otherwise processed before being transmitted to the frontend. The VGAs which provide a full-scale signal could be omitted andreplaced with digital processing element(s) that compensate for noise.The same LO reference need not be used to generate both periodicfunctions for the up-conversion and down-conversion—fixed phase orspread spectrum techniques may be employed instead. The upsampling stepis optional and may not be appropriate in various systems. For the sakeof brevity, the remaining circuit elements will not be individuallyaddressed here but it should be understood that various modifications tothe particular topologies may be made within the context of thisdisclosure.

It will thus be seen that the objects set forth above, among those madeapparent from the preceding description, are efficiently attained and,because certain changes may be made in carrying out the above method andin the construction(s) set forth without departing from the spirit andscope of the invention, it is intended that all matter contained in theabove description and shown in the accompanying drawings shall beinterpreted as illustrative and not in a limiting sense.

It is also to be understood that the following claims are intended tocover all of the generic and specific features of the invention hereindescribed and all statements of the scope of the invention which, as amatter of language, might be said to fall therebetween.

1-49. (canceled)
 50. A method for digitizing a signal, comprising:receiving an input analog signal spanning an original frequency rangethat exceeds a bandwidth of a first digitizer and that exceeds abandwidth of a second digitizer; splitting the received input analogsignal into at least a first split signal and a second split signal;digitizing the first split signal using the first digitizer; convertinga frequency range of the second split signal in a first conversionoperation; digitizing the second split signal using the seconddigitizer; converting the frequency range of the second split signal ina second conversion operation; and combining the first split signal andthe second split signal mathematically to form a single output datastream that provides a substantially correct representation of thereceived input analog signal, wherein the single output data stream hasa bandwidth of frequency content that has been digitized using at leastthe first digitizer and the second digitizer, and wherein the bandwidthof the frequency content exceeds a bandwidth of the first digitizer andexceeds a bandwidth of the second digitizer.
 51. The method of claim 50,wherein splitting the received input analog signal includes splittingthe input analog signal with a 50 Ohm splitter.
 52. The method of claim50, wherein converting the frequency range of the second split signal inthe first conversion operation includes mixing the second split signalwith a predetermined periodic function with a predetermined frequency.53. The method of claim 50, wherein the predetermined periodic functionis a low-distortion sinusoid.
 54. The method of claim 50, furthercomprising band limiting the first split signal to a first predeterminedband range, wherein the first predetermined band range spans a frequencyrange smaller than the original frequency range.
 55. The method of claim54, further comprising band limiting the second split signal to a secondpredetermined band range, wherein the second predetermined band rangespans a frequency range smaller than the original frequency range. 56.The method of claim 55, wherein: converting the frequency range of thesecond split signal in the first conversion operation includes mixingthe second split signal with a predetermined periodic function with apredetermined frequency; and a frequency of the predetermined periodicfunction is at a low side of the second predetermined band range. 57.The method of claim 55, wherein: converting the frequency range of thesecond split signal in the first conversion operation includes mixingthe second split signal with a predetermined periodic function with apredetermined frequency; and a frequency of the predetermined periodicfunction is at a high side of the second predetermined band range. 58.The method of claim 50, further comprising, in processing of the secondsplit signal that follows conversion of the frequency range of thesecond split signal in the first conversion operation, passing thesecond split signal through an image reject filter.
 59. The method ofclaim 58, wherein the image reject filter comprises an intrinsicbandwidth of a digitized channel used for digitizing the second splitsignal with the second digitizer.
 60. The method of claim 50, wherein:the first split signal spans a frequency range smaller than the originalfrequency range; and the second split signal spans a frequency rangesmaller than the original frequency range.
 61. The method of claim 50,wherein: converting the frequency range of the second split signal inthe first conversion operation includes converting the frequency rangeof the second split signal down in frequency; and converting thefrequency range of the second split signal in the second conversionoperation includes converting the frequency range of the second splitsignal up in frequency.
 62. The method of claim 50, wherein: convertingthe frequency range of the second split signal in the first conversionoperation includes converting the frequency range of the second splitsignal away from an initial frequency range of the second split signal;and converting the frequency range of the second split signal in thesecond conversion operation includes converting the frequency range ofthe second split signal back in frequency to the initial frequency rangefrequency range of the second split signal.
 63. The method of claim 50,wherein: converting the frequency range of the second split signal inthe first conversion operation occurs in processing of the second splitsignal that is before digitizing using the second digitizer; andconverting the frequency range of the second split signal in the secondconversion operation occurs in processing of the second split signalthat follows digitizing using the second digitizer.
 64. The method ofclaim 50, wherein combining the first split signal and the second splitsignal mathematically occurs in processing of the first split signal andprocessing of the second split signal that follows digitizing of thefirst split signal and digitizing of the second split signal.